Rc-oscillator circuit

ABSTRACT

An RC oscillator circuit is described in which a charge current (IPOSC 1 ) is integrated in an integrator ( 1 ). In a comparator ( 7 ), an output voltage (VCAP) of the integrator is compared to a reference threshold (VTH). Depending on the comparison, a periodic signal is generated in a clock pulse generator ( 9 ). Furthermore, a reference generator ( 8 ) is provided, which generates the reference threshold (VTH) depending on the temperature and on the supply voltage of the entire circuit. In this way, the frequency dependence of the oscillator is largely compensated for as far as fluctuations in the supply voltage are concerned.

The present invention relates to an RC oscillator circuit.

In RC oscillators, the output frequency is determined by RC networks, inother words by resistors and capacitors.

An RC oscillator can for example be realised in that a capacitor ischarged with a constant current, and the voltage resulting above thecapacitor is compared to a threshold value or reference value. As soonas the voltage above the capacitor exceeds a predetermined threshold,the capacitor is rapidly discharged. Subsequently, renewed charging ofthe capacitor takes place. Thus it is primarily the capacitor's chargingtime constant which determines the clock pulse rate or frequency of theoutput signal of the arrangement. If necessary, the resulting sawtoothvoltage can be converted to form a square-wave signal.

In the described principle of an RC oscillator, there is however aproblem in that the comparator, which compares the capacitor voltagewith the reference voltage, is subject to substantial fluctuations intemperature. This results in an undesirable dependence of theoscillator's output frequency on the ambient temperature.

There is a further disadvantage in that usually the described oscillatorprinciple depends to a significant degree on the supply voltage.Accordingly, fluctuations in the supply voltage also lead to significantundesirable fluctuations in the output frequency of the oscillator.

It is thus the object of the present invention to create an RCoscillator circuit in which the dependence of the output frequency ontemperature and/or on the supply voltage is reduced.

According to the invention, this object is met by an RC oscillatorcircuit comprising:

-   -   a current generator for generating a charge current;    -   an integrator with an input, which is coupled to the current        generator, and with an output;    -   a comparator comprising a first input which is connected to the        output of the integrator, and comprising a second input for        supplying a reference threshold;    -   a clock pulse generator which is connected to an output of the        comparator; and    -   a reference generator, designed for generating the reference        threshold, depending on a supply voltage of the RC oscillator        circuit.

According to the principle proposed, one input of the comparator iscoupled to an integrator which integrates a charge current, while afurther input is coupled to a reference generator. In this arrangement,the reference generator provides the reference threshold depending onthe supply voltage of the entire RC oscillator circuit.

By adjusting the reference threshold in relation to the supply voltage,a compensated reference threshold is provided so that the frequencyemitted by the clock pulse generator of the oscillator remains largelyindependent of fluctuations in the supply voltage.

Preferably, the integrator comprises at least one capacitor which ischarged by the charge current provided by the current generator.

Furthermore, preferably, a discharge device is provided which dischargesthe capacitor as soon as the voltage above the capacitor rises above thereference threshold. To this effect, the discharge device is preferablycontrolled by the comparator and/or the clock pulse generator.

As an alternative, the integrator can comprise two capacitors which arealternately charged and discharged. In this case, preferably a dischargedevice is provided for each capacitor. The charge currents for bothcapacitors can be fed to the capacitors by a common current generator,for example by way of a current mirror each.

Advantageously, the discharge current or discharge currents, too, can beprovided by the current generator by way of still further currentmirrors. In this arrangement it must be ensured that the time constantsfor discharge are clearly below those for charging the capacitors, sothat, advantageously, quick discharge of the capacitors can be carriedout.

In order to achieve still further improvement in the compensation of thereference threshold in the reference generator, it is further preferredthat the reference generator be coupled to the integrator such that thereference threshold is generated depending on the actual voltage abovethe capacitor, of which there is at least one. To this purpose, theactual voltage above the capacitor, of which there is at least one, ispreferably first integrated.

Thus, the reference threshold is generated depending both on the supplyvoltage of the oscillator and on the voltage above the capacitor, ofwhich there is at least one.

Advantageously, dependence of the reference threshold on the supplyvoltage is achieved in that by means of a programmable voltage divider,the supply voltage is first divided down to a settable value and in thatthe signal obtained in this way is further processed in the referencegenerator for the purpose of generating the reference threshold.

According to the proposed improvement relating to generating thereference threshold depending on the voltage above the capacitor of theintegrator, effects of the ambient temperature and/or effects whichfluctuations in the supply voltage have on the charging time constant ofthe capacitor in the integrator can be taken into account andcompensated for without any problems. As a result of this,advantageously, a comparator can be used whose requirements concerningspeed and accuracy are quite modest.

Preferably, the reference generator is switched into the RC oscillatorcircuit such that a control system is created which controls thereference threshold depending on the actual circuit circumstances in theRC oscillator, so that any fluctuations in the supply voltage and/orvariations in the charging time constant in the integrator can be“controlled away”, i.e. compensated for.

Preferably, the reference generator comprises an integrating amplifier.Preferably, the integrating amplifier comprises an input which iscoupled to the integrator, and comprises an output for supplying thereference threshold depending on the integrated voltage above thecapacitor, of which there is at least one. This results in still moreprecise compensation of temperature fluctuations and supply voltagefluctuations as well as production-related fluctuations of the RCoscillator circuit.

Furthermore, the reference generator preferably comprises a differentialamplifier which, at its output, provides the reference thresholddepending on the difference between a voltage derived from the supplyvoltage, and the integrated voltage above the capacitor, of which thereis at least one.

Preferably, the current generator comprises a voltage divider.Preferably, the voltage divider is connected to supply potential on theinput side, and connected to a voltage converter/current converter onthe output side. The voltage converter/current converter provides thecharge current.

Advantageously, the voltage converter/current converter comprises aresistor.

Further details and advantageous embodiments of the proposed principleare the subject of the subordinate claims.

Below, the invention is explained in more detail by means of exemplaryembodiments, with reference to several drawings.

The following are shown:

FIG. 1 a functional block diagram of an embodiment of an RC oscillatorcircuit according to the proposed principle;

FIG. 2 an exemplary schematic circuit diagram of a current generator;

FIG. 3 an example of a schematic circuit diagram of the integrator ofFIG. 1;

FIG. 4 an exemplary embodiment of a circuit of the reference generatorshown in FIG. 1;

FIG. 5 a simplified schematic circuit diagram of the reference generatorshown in FIG. 4 for explaining the way it functions;

FIG. 6 a further schematic circuit diagram for explaining the way thereference generator shown in FIG. 4 functions;

FIG. 7 an example of a diagram of a charge curve of a capacitor in theintegrator;

FIG. 8 an embodiment of a shift register for application in the clockpulse generator shown in FIG. 1;

FIG. 9 exemplary output signals of the clock pulse generator shown inFIG. 1;

FIG. 10 a D-flip-flop by means of a schematic circuit diagram forapplication in the clock pulse generator shown in FIG. 1; and

FIG. 11 the variation in time of selected signals for explaining the waythe proposed principle functions, with reference to the embodiment shownin FIG. 1.

FIG. 1 shows an RC oscillator circuit with reference to an embodimentaccording to the proposed principle. Provided is an integrator 1 with afirst input 2 for supplying a charge current IPOSC1, and an input 3 forsupplying a discharge current IPOSC2. Charge current and dischargecurrent are provided by a current generator (not shown in FIG. 1). Theintegrator 1 comprises two capacitors which are alternately charged anddischarged. The integrator 1 comprises three outputs 4, 5, 6. At output4, a capacitor voltage in the form of a scanning signal is provided,while the respective actual current on the two capacitors is present atoutputs 5 and 6. Output 4 is connected to a first input of a comparator7. A second input of the comparator 7 is connected to an output of areference generator 8. The reference generator 8 provides a referencethreshold VTH. An output of the comparator 7 is connected to an input ofa clock pulse generator 9 which in the present embodiment supplies clockpulse signals to three outputs 10, 11, 12, depending on the comparisonbetween the capacitor voltage VCAP and the reference threshold VTH inthe comparator 7. The clock pulse signals at the outputs 10, 11, 12 areindicated by the reference characters OSC1, OSC2, OSC3.

By way of a control bus 13, the clock pulse generator 9 is connected torespective control inputs of the integrator 1 and of the referencegenerator 8. The outputs 5, 6 of the integrator 1, at which outputs therespective capacitor voltages CAP1, CAP2 are present, are connected toinputs of the reference generator 8. A programmable voltage divider 14provides output voltages VTH1, VTH2, depending on a supply voltage ofthe entire RC oscillator circuit and depending on a programming word, atan input 15 of the voltage divider 14.

The frequency divider 14 comprises a 32-step voltage divider at whichthe second auxiliary voltage VTH2 is provided on the output side. Thevoltage divider is programmable with 5 bits by means of which the outputfrequency of the oscillator is determined. By programming the 5 bits,both the voltage at which the capacitors are charged, and the clockpulse period at the output are influenced. The auxiliary voltages VTH1,VTH2 are both derived from the supply voltage of the circuitarrangement.

Controlled by the clock pulse generator 9, two capacitors in theintegrator 1 are alternately charged and discharged. In the comparator7, the scanned capacitor voltage VCAP, which is ramp-shaped, is comparedto a reference threshold VTH. The reference threshold VTH among otherthings depends on the auxiliary voltages VTH1, VTH2. At every step ofthe capacitor voltage VCAP with the reference threshold VTH, thecomparator provides pulses. In the clock pulse generator 9, thefrequency of output signals OSC1, OSC2, OSC3 of the oscillator isdetermined from these pulses.

The reference threshold VTH depends on the capacitor voltages CAP1, CAP2in the integrator 1, with said capacitor voltages being integrated inthe reference generator 8. Furthermore, the reference threshold VTHdepends on the supply voltage of the circuit, with said supply voltagebeing divided down in the voltage divider 14 with a programmable dividerratio.

The control bus 13 controls the integrator 1 and the reference generator8 by means of a power-on-reset (POR) signal, a first changeover signalPRE1 and a second changeover signal PRE2. Further changeover signalsCHG1, CHG2 are used to drive the integrator 1. Additional changeoversignals CAL1, CAL2 are used to drive the reference generator 8.

Due to generation of the reference threshold which incorporatescompensation, the proposed principle makes it possible to compensate forboth fluctuations in temperature and fluctuations in the supply voltagewith regard to the output frequency of the oscillator. The frequency ofthe output signals OSC1, OSC2, OSC3 of the oscillator is thusadvantageously largely independent of the ambient temperature and of thesupply voltage of the circuit.

The defined clock pulse period is obtained in that capacitors arecharged with a constant current and in that the voltage ramp signal iscompared with the reference threshold. Charge currents and dischargecurrents are provided in a way that does not depend on the temperature.

The function of the individual circuit blocks and their advantageouscombined action is explained in more detail by way of examples withreference to the enclosed figures.

FIG. 2 shows an exemplary embodiment of a current generator whose inputs2, 3 are coupled to the integrator 1, with said current generator beingdesigned to generate a charge current IPOSC1 and to generate a dischargecurrent IPOSC2. The current generator shown in FIG. 2 comprises avoltage divider 16, 17 which comprises a series connection with tworesistors which are switched between the supply potential 18 and thereference potential 20. A tapping-off node between the resistors 16, 17is connected to the inverting input of an operational amplifier 19. Byway of its output, the operational amplifier 19 controls the gateconnection of a field effect transistor 21. By way of an externalresistor 22, the drain-connection of the transistor 21 is connected toreference potential 20 as well as to the non-inverting input of theoperational amplifier 19. The source connection of the transistor 21 isconnected to a multitude of current mirror circuits 23, 24, 25, 26, 27,28, 29 which provide practically any number of reference currents. Theoutput currents which can be tapped off at the current mirrors 23 to 29are temperature-independent.

Part of the supply voltage VDD at the corresponding connection 18 issupplied to the negative input of the transconductance amplifier 19. Thesame voltage is present at the non-inverting input of thetransconductance amplifier 19 and drops by way of the external resistor22. In order to minimise so-called on-chip resistors serially to theexternal resistor 22, the connection between the resistor connection andthe plus connection of the amplifier 19 is implemented externally. Theoutput of the amplifier 19 controls the gate connection of a p-channelMOS field effect transistor. A negligible current is absorbed by thepositive input connection of the resistor so that the resulting currentwhich flows through the controlled section of the transistor 21 and thusalso through the external resistor 22 is according to the equation:$I_{REXT} = {V_{DD}\frac{R_{A}}{R_{A} + R_{B}}\frac{1}{R_{EXT}}}$with R_(A), R_(B) denoting the resistance values of the voltage divider16, 17, with V_(DD) denoting the voltage at the supply potentialconnection 18, and with R_(EXT) denoting the value of the externalresistor 22.

If the mirror ratios of the current mirrors are included, then thegeneral current which is generated by the current generator and which isindependent of the temperature is:$I = {V_{DD}\frac{R_{A}}{R_{A} + R_{B}}{\frac{1}{R_{EXT}} \cdot k}}$With this constant current, a capacitance is charged.

FIG. 3 shows an embodiment of an integrator 1. Two capacitors C1, C2have been provided, each of which is connected via a switch 30, 31 to aconnection with the input 2 of the integrator, and with a furtherconnection each being connected to the supply potential connection 18.By way of a current mirror 32 and a further switch 33, 34 each, theinput 3 is also connected to one each of the capacitors C1, C2. Aconnection each of capacitors C1, C2 is connected to an output 5, 6 eachof the integrator, which are coupled to the reference generator 8. Theoutput 4 of the integrator 1, to which the comparator 7 is connected, isconnected to a capacitor C1, C2 each, by way of a switch 35, 36 each.

The two capacitors C1 and C2 are alternately charged and discharged. Theswitches 35, 36 are activated with control signals CHG1′, CHG2′ suchthat the output 4 is always connected to that capacitor C1, C2 which isbeing charged at the time. The changeover signals CHG1 and CHG2, whichare provided by the clock pulse generator 9, select the charge phase ofthe capacitors C1, C2 by driving the switches 30, 31. The dischargesignals PRE1 and PRE2, which drive the switches 33, 34, select thedischarge phase of the respective capacitor.

In this example, the capacitors C1 and C2 are designed as respectivegate capacitances of MOSFET, Metal Oxide Semiconductor Field EffectTransistor structures. In this arrangement, for each metal option in theintegrated circuit several capacitors switched in parallel can beselected to change the clock pulse frequency.

The switches 30, 31, 33, 34, 35, 36 are transmission gates, eachcomprising an n-channel and a p-channel MOS field effect transistor. Theswitches 30, 31, 33, 34, 35, 36 are in the closed state if therespective control signal is High in relation to the n-channeltransistor switch, and is Low in relation to the p-channel transistorswitch.

The constant currents IPOSC1 and IPOSC2 at the inputs 2, 3 are used forcharging the capacitor selected by the respective switch, or fordischarging said capacitor. The current mirror 32 has a mirror ratio of5:1 so that discharge of the capacitors C1, C2 takes place very rapidlyin relation to the charge procedure. The charge is completely withdrawnfrom the respective capacitor before the next charge phase commences.The voltage during the discharge phase remains linear until the currentmirror transistor on the output side leaves saturation, at which pointthe voltage becomes exponential.

When starting the entire oscillator circuit by means of the controlsignal POR, the capacitor C2 is charged to the differential voltage ofthe supply voltage VDD minus the auxiliary voltage VTH2 of the voltagedivider 14, and the capacitor C1 is charged to VDD.

A clock pulse period at the output of the clock pulse generator 9 iscomposed of four clock pulse phases. During the first clock pulse phasethe capacitor C1 is charged and the voltage at the capacitor C2 isconstant. As soon as the voltage above the capacitor C1 intersects thefirst auxiliary threshold VTH1, the second clock pulse phase commences.During the second clock pulse phase, C1 continues to be charged, howeverC2 is already being discharged. The third clock pulse phase starts whenthe voltage above the capacitor C1 reaches the second auxiliarythreshold VTH2. During the third clock pulse phase, the charge on thecapacitor C1 remains constant, while C2 is being charged. When thevoltage above the capacitor C2 intersects the auxiliary threshold VTH1,the fourth and last clock pulse phase commences. In the fourth clockpulse phase, C2 continues to be charged, however C1 is being discharged.The output of the reference generator 8, i.e. the output voltage VTH,periodically assumes the auxiliary threshold VTH1, provided by thedivider 14, and a voltage value which results from correcting thethreshold VTH2. The second auxiliary threshold VTH2 is adjusted in eachhalf-period according to the error which the comparator 7 makes duringcommuting of the preceding half-period. The first auxiliary thresholdVTH1 is constant and is used to split each half-period into two furthersub-periods. In each instance, the error is estimated during the firstphase of each half-period.

FIG. 4 shows an exemplary embodiment of the reference generator 8 whichis shown in FIG. 1. At the output of the reference generator 8, thereference threshold VTH is provided which is compared in the comparator7 to the respective capacitor voltage signal in the charge phase.

The reference generator 8 comprises a differential amplifier 37. For thepurpose of supplying the auxiliary voltage VTH2, the non-inverting inputof the differential amplifier 37 is connected to the output of thevoltage divider 14. The inverting input of the differential amplifier 37is connected to an input node K by way of a series capacitor CA, and isconnected to the output of the differential amplifier 37 by way of afeedback capacitor CB. The series capacitor CA can be short-circuited byway of two series switches 38, 39 connected in parallel, with saidseries switches being driven by the clock pulse generator 9 by means ofdischarge signals PRE1, PRE2. The input node K is connected to theoutput 5 of the integrator 1 by way of a switch 40, and to the output 6of the integrator 1 by way of a switch 41, which in each case supply thevoltage CAP1, CAP2 by way of capacitors C1, C2. The switches 40, 41 arecontrolled by changeover signals CAL1, CAL2 from the clock pulsegenerator 9. The output of the differential amplifier 37 is connected tothe output of the reference generator 8 by way of two switches 42, 43connected in parallel, with said switches 42, 43 being controlled bydischarge signals PRE1′, PRE2′. Two further switches 44, 45, which arealso connected in parallel, connect the voltage divider 14 via theconnection for supplying the auxiliary voltage VTH1 to the output forproviding the switching threshold VTH of the reference generator 8.

Below, the function of the circuit shown in FIG. 4 is explained,starting from an ideal case. In an ideal case, in which an idealcomparator is used, the value of the reference threshold VTH is equal tothe value of the auxiliary thresholds VTH1 and VTH2. In a real casehowever, the comparator switches at the point where the ramp voltageattains the value of the second auxiliary threshold VTH2 plus an errorvoltage VERR. An error in the clock pulse period results, which error iscalculated from the product of the voltage error VERR and the quotientfrom the capacity C and the current I, where C represents the chargingcapacity and I represents the charge current. In the next half-period,VTH first assumes the value of the first auxiliary threshold VTH1.Subsequently, the reference threshold VTH is arrived at, according tothe equation: ${VTH} = {{{VTH}\quad 2} - {\frac{CA}{CB} \cdot {VERR}}}$

CA designates the capacity of the series capacitor CA, and CB designatesthe capacity of the feedback capacitor CB of FIG. 4.

This results in an error in the clock pulse period of$\left( {1 - \frac{CA}{CB}} \right) \cdot {VERR} \cdot \frac{C}{I}$

If the comparator error remains constant, according to the principleproposed, the error of the clock pulse period is reduced to the valuerepresented by the following equation, after going through a number N ofsteps: $\left( {1 - \frac{CA}{CB}} \right)^{N}{VERR}\frac{C}{I}$

FIGS. 5 and 6 show simplified schematic circuit diagrams starting withFIG. 4, by means of which the function of the circuit in FIG. 4 isexplained. In this process, it should first be assumed that in a firsttime interval T between the points in time T0 and T1, T0<T<T1, thevoltage V1 above the feedback capacitor CB should equal 0. In the firsttime interval, the resulting ratios areV0=0, CAL=0,with V0 representing the voltage above the series capacitor CA, and withCAL representing the switching signal for the switch, shown in FIG. 6,which is connected in series with the capacitor CA.

A second time interval T1<T<T2 is examined. The voltage above thereplacement voltage source CAP shown in FIG. 6 is CAP=VTH2+VERR. Fromthis follows:V0=VTH2+VERR−VTH2=VERR.

The following results:Q0=CA×V0=CA×VERR.

The following applies: If the switch, controlled by signal CAL closesand the parallel switch to the series capacitor CA (the switch signalbeing designated PRE) is opened, a current which flows through CAsupplies the charge Q0. The same current also flows through CB, as shownin FIG. 5, because the input of the amplifier is highly resistive. Fromthis follows: Q  0 = Q  1, so  that${VI}^{\prime} = {\frac{Q\quad 0}{CB} = {\frac{CA}{CB} \cdot {VERR}}}$and ${VTH}^{\prime} = {{{VTH}\quad 2} - {\frac{CA}{CB} \cdot {VERR}}}$

In a third time interval T2<T<T3 the following applies: The switch,controlled by signal CAL is open, the capacitor CB receives the chargeand its voltage V1′, while V0 above CA equals 0.

In a fourth interval T3<T<T4 the following applies: $\begin{matrix}{{CAP} = {{VTH}^{\prime} + {VERR}}} \\{= {{{VTH}\quad 2} - {\frac{CA}{CB} \cdot {VERR}} + {VERR}}} \\{= {{{VTH}\quad 2} + {{VERR}\left( {1 - \frac{CA}{CB}} \right)}}}\end{matrix}$ ${V\quad 0} = {{VERR}\left( {1 - \frac{CA}{CB}} \right)}$${Q\quad 0} = {{CA} \cdot {{VERR}\left( {1 - \frac{CA}{CB}} \right)}}$$\begin{matrix}{{VI}^{''} = {{VI}^{\prime} + \frac{Q\quad 0}{CB}}} \\{= {{\frac{CA}{CB}{VERR}} + {\frac{CA}{CB}{{VERR} \cdot \left( {1 - \frac{CA}{CB}} \right)}}}} \\{= {{VERR}\left\lbrack {{2\frac{CA}{CB}} - \left( \frac{CA}{CB} \right)^{2}} \right\rbrack}}\end{matrix}$${VTH}^{''} = {{{VTH}\quad 2} - {{VERR} \cdot \left\lbrack {{2\frac{CA}{CB}} - \left( \frac{CA}{CB} \right)^{2}} \right\rbrack}}$

In a fifth interval T4<T<T5, the same applies as in interval 3, apartfrom the fact that CB is supplied with the voltage V1″.

In a sixth interval, the following applies: T5<T<T6 $\begin{matrix}{{CAP} = {{VTH}^{''} + {VERR}}} \\{= {{{VTH}\quad 2} - {{VERR}\left\lbrack {{- 1} + {2 \cdot \frac{CA}{CB}} - \left( \frac{CA}{CB} \right)^{2}} \right\rbrack}}} \\{= {{{VTH}\quad 2} + {{VERR}\left( {1 - \frac{CA}{CB}} \right)}^{2}}}\end{matrix}$${V\quad 0} = {{VERR}\left( {1 - \frac{CA}{CB}} \right)}^{2}$${Q\quad 0} = {{CA} \cdot {{VERR}\left( {1 - \frac{CA}{CB}} \right)}^{2}}$$\begin{matrix}{{VI}^{''} = {{VI}^{\prime} + \frac{Q\quad 0}{CB}}} \\{= {{{VERR}\left\lbrack {{2\frac{CA}{CB}} - \left( \frac{CA}{CB} \right)^{2}} \right\rbrack} + {\frac{CA}{CB}{{VERR}\left( {1 - \frac{CA}{CB}} \right)}^{2}}}} \\{= {{VERR}\left\lbrack {{3\frac{CA}{CB}} - {3\left( \frac{CA}{CB} \right)^{2}} + \left( \frac{CA}{CB} \right)^{3}} \right\rbrack}}\end{matrix}$${{VTH}^{''\prime} = {{{VTH}\quad 2} - {{VERR}\left\lbrack {{3\frac{CA}{CB}} - {3\left( \frac{CA}{CB} \right)^{2}} + \left( \frac{CA}{CB} \right)^{3}} \right\rbrack}}},{{etc}.}$

By means of a diagram, FIG. 7 shows the voltage over time, at which thecapacitors C1, C2 shown in FIG. 3 are charged. Note that with thevoltage divider 14, which comprises a multiplexer, it is not onlypossible to change the voltage at which the capacitors are charged, butalso to change the clock pulse period.

FIG. 8 shows a shift register which is encompassed by the clock pulsegenerator 9. The shift register comprises four register cells 46, 47,48, 49 which are interconnected in a ring-shaped manner, with eachregister cell being clock pulsed with the output signal of thecomparator 7. For each clock pulse period, four clock pulse phases areprovided. Each clock pulse phase starts with a switch pulse of thecomparator 7. In an initialisation state, the register cells 46 and 47are precharged with a logical 1, while the register cells 48 and 49 areprecharged with a logical 0. Consequently, the starting state of theshift register is 1100. The internal state of the shift register changeswith each pulse on the COMP signal of the comparator.

Consequently, the sequence of states of the shift register is:

-   1100-   0110-   0011-   1001-   1100 etc.

The internal states of the shift register 46, 47, 48, 49 generate fourauxiliary signals of which the periodic square-wave signal OSC1 at theoutput of the clock pulse generator 9 is composed. The further outputsignals OSC2 and OSC3 are arrived at by dividing the output signal OSC1by 2, or by 32 respectively. FIG. 9 shows an example of the clock pulsegradients of the output signals OSC1 and OSC2.

It is important to note that all auxiliary signals for controlling thefour phases of a clock pulse period are generated from the internalstates of the shift register 46, 47, 48, 49 of FIG. 8 by logicoperations.

FIG. 10 is an example of a schematic circuit diagram of a positive,flank-triggered register by means of a D-flip-flop with master-slavestructure.

For a better understanding of the function of the circuits shown inFIGS. 1 to 4, FIG. 11 shows the time gradients of selected signals. Theswitching signals CHG1 and CHG2 are used for controlling switches 30 and31 in the integrator 1 of FIG. 3, with said switching signals beingprovided by the clock pulse generator 9. The signals PRE1 and PRE2 areused to switch the discharge switches 33 and 34. The switching signalsCALL and CAL2 are used to control the scanning switches 40 and 41 in thereference generator 8. CAP1 and CAP2 designate the capacitor voltages atthe outputs 5, 6 of the integrator 1. VCAP designates the voltagegradient at the output 4 of the integrator 1, wherein in every case thecharging phases of the capacitors C1, C2 are switched to the output. VTHshows the signal gradient of the reference threshold which is providedby the reference generator 8. At the beginning of each of the fourphases of a clock pulse period, the output signal COMP shows a pulse.OSC designates the output signal OSC1 of the clock pulse generator 9. Itis shown that a clock pulse period of the RC oscillator comprisesexactly four clock pulse phases.

For example a power-on-reset signal, a scanning signal for 5 programmingbits, as well as the 5 programming bits for specifying the desiredoutput frequency are provided as input signals of the proposedintegrated RC oscillator. Three clock pulse frequencies, four n-type andtwo p-type currents are the output signals. Furthermore, a connectionpin for connection to an external resistor is provided, which providesthe temperature-independent current.

Of course it is within the scope of the invention to also use otherembodiments of the oscillator circuit, which embodiments differ fromthose shown.

LIST OF REFERENCE CHARACTERS

-   1 Integrator-   2 Charge current input-   3 Discharge current input-   4 Output-   5 Auxiliary output-   6 Auxiliary output-   7 Comparator-   8 Reference generator-   9 Clock pulse generator-   10 Output-   11 Output-   12 Output-   13 Control bus-   14 Voltage divider-   15 Input-   16 Resistor-   17 Resistor-   18 Supply potential connection-   19 Operational amplifier-   20 Reference potential connection-   21 Transistor-   22 Resistor-   23 to 29 Current mirror-   30 Switch-   31 Switch-   32 Current mirror-   33 to 36 Switch-   37 Differential amplifier-   38 to 45 Switch-   46 to 49 Register cell

1-10. (canceled)
 11. An RC oscillator circuit, comprising: a currentgenerator configured to generate a charge current; an integrator havingan input and an output, the input being connected to the currentgenerator; a comparator having a first input, a second input, and anoutput, the first input being connected to the output of the integratorand the second input being configured to supply a reference threshold; aclock pulse generator connected to the output of the comparator; and areference generator configured to generate the reference threshold basedon a supply voltage of the RC oscillator circuit, wherein the integratorcomprises a first capacitor and a second capacitor which are alternatelycharged and discharged.
 12. An RC oscillator circuit, comprising: acurrent generator configured to generate a charge current; an integratorhaving an input and an output, the input being connected to the currentgenerator, the integrator comprising at least one capacitor; acomparator having a first input, a second input, and an output, the afirst input being connected to the output of the integrator and thesecond input being configured to supply a reference threshold; a clockpulse generator connected to the output of the comparator; and areference generator coupled to the integrator, the reference generatorbeing configured to generate the reference threshold based on a supplyvoltage of the RC oscillator circuit, and a voltage at a node above thecapacitor.
 13. The RC oscillator circuit of claim 11, wherein theintegrator further comprises a discharge device configured to dischargeat least one of the first capacitor and the second capacitor.
 14. The RCoscillator circuit of claim 11, wherein the reference generatorcomprises an integrating amplifier, the integrating amplifier having aninput and an output, the input being coupled to the integrator and theoutput being configured to supply the reference threshold based on anintegrated voltage at a node above at least one of the first capacitorand the second capacitor.
 15. The RC oscillator circuit of claim 12,wherein the reference generator further comprises a differentialamplifier configured to generate the reference threshold based on adifference between a voltage derived from the supply voltage and thevoltage at the node above the at least one of the first capacitor andthe second capacitor.
 16. The RC oscillator circuit of claim 11, whereinthe current generator comprises a voltage divider having an inputconnected to a supply potential connection and an output connected to avoltage-to-current converter.
 17. The RC oscillator circuit of claim 16,wherein the voltage-to-current converter comprises a resistor.
 18. TheRC oscillator circuit of claim 11, wherein the current generator iscoupled to the integrator by at least one current mirror.
 19. The RCoscillator circuit of claim 12, where the integrator further comprises adischarge device configured to discharge the at least one capacitor. 20.The RC oscillator circuit of claim 12, wherein the reference generatorcomprises an integrating amplifier, the integrating amplifier having aninput and an output, the input being coupled to the integrator and theoutput being configured to supply the reference threshold based on anintegrated voltage at a node above the at least one capacitor.
 21. TheRC oscillator circuit of claim 20, wherein the reference generatorcomprises a differential amplifier configured to generate the referencethreshold based on a difference between a voltage derived from thesupply voltage and the voltage at the node above the at least onecapacitor.
 22. The RC oscillator circuit according of claim 12, whereinthe current generator comprises a voltage divider having input connectedto a supply potential connection and an output connected to avoltage-to-current converter.
 23. The RC oscillator circuit of claim 22,wherein the voltage-to-current converter comprises a resistor.
 24. TheRC oscillator circuit of claim 12, wherein the current generator iscoupled to the integrator by at least one current mirror.